Modulator/transmitter apparatus and method

ABSTRACT

Method and apparatus for providing an amplified modulated radio frequency signal, the method comprising the steps of providing a radio frequency (RF) oscillatory signal, generating from the RF oscillatory signal and an input signal a pair of phase modulated phase conjugated signals, and summing said pair of signals to provide the desired modulated RF signal output. A form of the invention is applied to digital modulation using in-phase (I) and quadrature (Q) input signals. Each of the I and Q signals is processed in a respective channel, with the channel outputs being summed. The channels share an RF oscillatory signal, which is phase shifted by 90 degrees within the Q channel.

This invention relates to radio frequency modulation and transmission.The invention is particularly, but not exclusively, useful in handlingdigital radio signals such as those used in mobile phone systems.

One problem for all mobile phone manufacturers is the inability of theindustry and the national standardisation authorities to define globalregulations regarding the modulation schemes for mobile communications.As a consequence, to date a single transmitter type cannot be truly“global”. Therefore, the current design philosophy is to build mobilephones that contain multiple transmitters, whereby only the part thatcomplies with the local modulation scheme is activated. The alternativeis to have mobiles that work only in a limited geographic region. Thishowever is costly and results in bulky and/or heavy mobiles and/orunsatisfied customers who find their handsets unusable when goingaboard. It also means that whenever modulation schemes have to bechanged to incorporate technical changes, e.g. during the transitionfrom the 2G to 3G networks, old equipment becomes obsolete.

Consequently, the industry has tried for quite some time to develop“software radios”. A software radio contains only standard componentsthat can reconfigure themselves on request of the controlling software,hence adapt the transmitter to comply with the relevant technicalrequirements. The problem is particularly acute at the modulator, upconverter, amplification chain. However, attempts so far have not led toany major breakthrough because they were based on modifications to aclassic superheterodyne transmitter, i.e. keep the traditionaltransmitter topology (modulation specific baseband modulators followedby up converters followed by the linear power amplifiers, necessary topreserve digital data modulation characteristic integrity),e.g.[1].

The present invention resides in a method for providing an amplifiedmodulated radio frequency signal, the method comprising:

-   -   providing a radio frequency (RF) oscillatory signal;    -   generating from the RF oscillatory signal and an input signal a        pair of phase modulated phase conjugated signals; and    -   summing said pair of signals to provide the desired modulated RF        signal output.

Said pair of signals may be produced by modulating said RF oscillatorysignal with the input signal to provide a modulated RF signal, andcombining the modulated RF signal with a multiple of the RF oscillatorysignal to provide a composite signal.

Preferably, each of the modulated RF signal and the composite RF signalis separately amplified before being combined with the other. For bestDC to RF efficiency amplification may be non-linear.

Preferably, said multiple is 2, and may be produced from the oscillatorysignal by a frequency multiplier, or by use of a harmonic mixer.

In one form of the invention, the input signal is an analogue signal andsaid modulation is an analogue phase shift. The result of the summationis an amplitude modulated RF signal. Optionally, the amplitude modulatedsignal may be further summed with said oscillatory RF signal to providea frequency modulated RF signal.

More significantly, however, another form of the invention is appliedwhere the input signal comprises in-phase (I) and quadrature (Q) inputsignals, which are digitally modulated. Each of the I and Q signals isprocessed in a respective channel by the foregoing method, the channeloutputs being summed. The channels share an oscillatory RF signal, whichis phase shifted by 90 degrees within the Q channel.

From another aspect, the invention provides a modulator for generating amodulated RF signal, the modulator comprising:

-   -   a local oscillator generating an RF oscillatory signal;    -   means for producing, from said RP oscillatory signal and an        input signal, a pair of phase modulated phase conjugated        signals; and    -   a summing circuit receiving said pair of signals as inputs to        generate a modulated RF output signal as its output.

Said means for producing a pair of phase modulated phase conjugatedsignals typically comprises a modulator arranged to modulate said RFoscillatory signal with the input signal to provide a modulated RFsignal, and a mixer connected to mix said modulated RF signal with amultiple of the RF oscillatory signal to provide a composite signal.

Preferably, a first amplifier is connected between the mixer output anda respective input of a first summer, and a second amplifier isconnected between the modulator output and a respective input of a firstsummer.

The mixer may be a fundamental mixer and may receive the localoscillator output via a multiplier circuit, typically a doubler.Alternatively, the mixer may be a harmonic mixer connected directly tothe local oscillator output.

In one form of the invention, for use with an analogue input signal, themodulator is an analogue phase shifter and the output of the summingcircuit is an amplitude modulated RF signal. The modulator may include afurther summer which sums the output of the first summer with the localoscillator signal to provide a frequency modulated output signal.

More typically, however, the input signal comprises in-phase (I) andquadrature (Q) input signals, which are digitally modulated. In thiscase, the modulator comprises an I channel and a Q channel each asdefined above and sharing a common local oscillator, the outputs of thesummers of the I and Q channels being combined by a third summer toprovide the modulated RF output signal.

Each channel may comprise a mixer. The mixers may be fundamental mixerssupplied with a frequency doubled local oscillator signal. Each channelmay have its own frequency doubler, with that in the Q channel receivingthe local oscillator output phase shifted by 90 degrees. Alternatively,both mixers may be supplied by a single frequency doubler, and thesignal from the local oscillator to the Q channel modulator be phaseshifted by 90 degrees. In a further option, the mixers may be harmonicmixers, in which case frequency doublers are not required.

In another embodiment, the means for producing the pair of phasemodulated phase conjugated signals are constituted by a see-sawmodulator in each channel.

The see-saw modulator may suitably use shorted lines to create thenecessary phase shifts for a phase modulated signal, and the lines maybe selectively shorted electronically, for example by switching of PINdiodes.

In the present invention, in contrast to the classic solutions,modulation is not performed in the baseband, but directly at the RFfrequency, through the application of simple phase shifters rather thancomplicated IQ modulators. The in-phase and quadrature part of thesignal (synthesised using phase conjugation vector summation) aremodulated and amplified separately, and finally combined, to form theactual IQ modulation specific signal. This results in several advantagesover the classic approach, which will be detailed below.

Embodiments of the present invention will now be described, by way ofexample only, with reference to the drawings, in which:

FIG. 1 is a schematic of the system topology of one embodiment of theinvention;

FIGS. 2 to 5 are phase diagrams illustrating phase states at variouspoints in FIG. 1;

FIG. 6 is a schematic of a second embodiment, also indicating amodification thereof;

FIG. 7 shows a further embodiment;

FIGS. 8 to 12 are phase diagrams illustrating the system of FIG. 1operating in other modulation schemes;

FIG. 13 shows the application of the invention in embodiments forproducing AM and FM signals;

FIG. 14 is a phase diagram showing the output of a simulation of theembodiment of FIG. 1 using amplifiers operating in linear mode;

FIGS. 14 a and 14 b show measured phase states for the arrangement ofFIG. 1;

FIG. 15 shows the output of a similar simulation but with non-linearamplifiers; and

FIG. 16 shows harmonic content at various points in the simulatedcircuit of FIG. 15.

1.1 OPERATING PRINCIPLE

FIG. 1 shows the overall system topology of the suggested novelarchitecture used to produce adaptive digital modulation and RFtransmission with highly efficient DC to RF amplification. The twoconventional baseband input signals. I and Q, are fed to the system, andthe modulated and amplified output signal, called RF, is taken from thesystem. The system requires a single local oscillator (LO), which can bederived in the usual way, e.g. from a phase locked loop frequencysynthesiser.

The purpose of the transmitter shown in FIG. 1 is to produce anamplified RF output signal with a pre-specified digital modulationscheme applied to it. For the purpose of introductory explanation of theworking principle, it is assumed that quadrature phase shift keyedmodulation, QPSK [11], of the baseband signals is to be synthesised andtransmitted on an RF carrier.

1.1.1 The Phase Modulation Stage

The two baseband input signals I and Q, here assumed to be digital, areeach fed to the phase setting control port of a phase modulator,designated here PM_(I) and PM_(Q), respectively, FIG. 1. The LO signal,at the requisite carrier frequency, is fed to the input ports of thephase modulators and a phase shift is applied to it as determined by thelogic state of the I,Q bit pattern applied. A 90° phase shift isintroduced in the quadrature channel with respect to the in-phasechannel. Consequently, the output signal of each of the phase modulatorswill vary between two discrete phase states.

To achieve QPSK, one possible set of output phase states is 45° and 135°for signal 1, and 135° and −45° for signal 2, respectively, FIG. 1. Thetwo signals are shown in FIG. 2. These phase states are obtained byusing simple software or analogue to digital hardware to select theappropriate phase shifts through phase modulators PM_(I), PM_(Q).

Signals 1 and 2 are then split in to equal strength signal paths, ofwhich one is directly amplified. This results in the signals 1′ and 2′,receptively. The other signals are each fed to separate RF balancedmixers, called MIX_(I) and MIX_(Q).

1.1.2 The Phase Conjugation Stage

The LO signals for mixers MIX_(I), MIX_(Q) are derived from the originalcarrier signal by frequency doubling it. Hence, the LO signal applied tothe mixers is phase locked to the initial carrier signal, and willintroduce a constant phase shift into the IF output signal of thesemixers. The LO signal for mixer MIX_(Q) is derived from the initialcarrier signal after a 90° phase shift has been applied. This introducedphase shift is then doubled to become 180° at mixer MIX_(Q).

Although not a pre-requisite for operation, double balanced mixers areused for MIX_(I) and MIX_(Q), since these naturally suppress unwantedleakage signals. The difference signal will have a frequency equal tothe initial carrier frequency, because the LO signal is exactly twicethe RF signal. Furthermore, it is an inherent property of the differencesignal that its phase is conjugated with respect to the RF input signal.It is this property that allows direct synthesis of the digitalmodulation schemes.

Hence, additional phase states of the IF output signals after thismixing process occurs become available. These result from the fact thatthe phase of the lower sideband after the mixing process is conjugatedwith respect to the input signal, i.e a phase of φ(t) for the lowersideband output signal. For example, if the phase of the LO signal isassumed to be 0°, and the phase of the RF signal say from MIX_(I) inFIG. 1 is φ, then Signal 3 at the output from MIX_(I) will becos(2ω_(LO) t+0°)cos(ω_(LO) t+φ)i.e ½ cos(3ω_(LO) t+φ) and ½ cos(ω_(LO) t−φ)

The up-converted term can be suppressed by the amplifier frequencyresponse, while the down converted term contains the wanted conjugatephase of the RF phase shifted signal. Hence, the phase states in FIG. 2become phase conjugated to those in FIG. 3, i.e. −45° and −135° forsignal 1, and 45° and −135° for signal 2. Note that in the latter case,the signal has an additional phase shift of 180° due to the introducedLO phase shift. Consequently, the phase states of signals 3 and 4 are asshown in FIG. 3.

Finally, the phase conjugated signals are amplified, resulting insignals 3′ and 4′, respectively, these are then combined after suitablepower factor correction with a non-conjugated signals 1′ and 2′,respectively, see section 1.1.3. Thus, an RF output signal which has theappropriate modulation scheme applied, has been amplified and set fortransmission. Note here that, since the I, Q bit patterns have beenconverted to PM modulated constant amplitude signals, highly efficientnon-linear amplification can be used without deteriorating theproperties of the modulated signal being transmitted.

1.1.3 The Combination Stages

In the first step, signal 1′ is vectorially combined with signal 3′, andsignal 2′ with signal 4′, the resulting output signals, 5 and 6, areshown in FIG. 4. It can be seen that the two signals are now inquadrature with respect to each other. When these signals arevectorially combined in the final combination stage, we achieve thewanted QPSK signal, FIG. 5.

1.1.4 Properties of Amplifier

It is important to note here that not only have we created a vectormodulator based on simple phase shifters using a phase conjugationphilosophy, but also as an additional benefit we have created asituation whereby we can use highly non-linear (i.e DC to RF efficient)amplifiers, because the RF signal is of constant amplitude duringamplification.

The principle of converting an amplitude modulated signal into a phasemodulated one during amplification was first reported by Cox in 1974[2]. However, Cox generated his constant amplitude signal only after theup conversion, on the RF frequency and within the amplifier. His ideahad therefore the severe disadvantage of resulting in a very high countof very complex and technically challenging components used to generatethe AM to PM conversion requisite for the technique to work. Inaddition, adaptive modulation as a wanted artefact of the AM to PMconversion was not addressed. Our improvements, to firstly generate theconstant amplitude signals during the modulator process, and second, theuse of quadrature signals to achieve any given digital modulation schemewithout changes to the hardware, results in a far more flexible andsimpler system.

1.2 ALTERNATIVE SYSTEM TOPOLOGIES 1.2.1 System for use withFundamental/Harmonic Mixers

The system topology as shown in FIG. 1 was chosen, because the mixersMIX_(I) and MIX_(Q) ultimately can be replaced by harmonic mixers [3],making the need for frequency doublers redundant.

However, should harmonic mixers not be chosen due to their inherentlylower conversion gain, another simplification to reduce component countfor the system is shown in FIG. 6. Here, the required phase shift tobring the Q channel in quadrature to the I channel is not added to theLO signal at the mixer MIX_(Q), but directly inserted on the carrierbefore it is fed to the phase modulator, PM_(Q). It could also beinserted at any other point in the Q path, but this position shown waschosen because there is no modulation present, which means that thephase modulator does not have to exhibit significant bandwidthperformance other than required for the LO in a multichannel situation.

A further modification could be to replace power factor correctedcombiner 7 with a spatial power combiner, i.e by feeding to a multiportantenna or an array of antennas [4].

1.2.2 System Using See-Saw Modulator

A very simple way to create phase conjugated phase modulated signals isby using a See-Saw modulator. The See-Saw modulator uses the reflectionsat shorted lines to create the necessary phase shifts for a discretephase modulated signal. The essential property of the See-Saw modulatoris that the output from port 1, in FIG. 7 is the phase conjugate of thatcoming from port 3. The different phase shifts from different phasestates are achieved by shorting the lines at different discrete pointsusing PIN diodes as switches. The actual combination of PIN open andshort circuit states needed to fulfil the phase states necessary foreach digital modulation type, Section 2, would be determined by simplehardware or software logic. However, it is obvious that therefore onlydiscrete phase states can be realised, and the number of possible phasestates is predetermined by the number of PIN diodes integrated into thesystem, and can no longer be changed after the system has beenmanufactured. Also, there is of course no analogue modulation possible.A detailed description of the modulator can be found in [5].

Using the See-Saw modulator to directly modulate the RF carrier, leadsto significant simplifications in the system, as there is no longer aneed to create an LO signal of twice the RF frequency. The modifiedsystem can be seen in FIG. 7.

The main advantage of this system over the one introduced in Section 1.1is the very simple set-up, which allows to adopt any given modulationscheme just by switching PIN diodes according to a preset scheme, whichcould for example be stored in a look-up table. Also, having eliminatedthe need for the LO signal of twice the RF frequency, the requirementfor one, or two frequency doublers, respectively, and for the twobalanced mixers has been removed, significantly simplifying the system.

2 UTILITY OF SYSTEM WITH OTHER MODULATION SCHEMES 2.1 BPSK [11] 2.1.1Classic Phase State Arrangement

For classic BPSK, only the I signal is used, the Q-Port is not required,but included here for uniformity of presentation with more complexmodulation schemes. In order to synthesise the desired output phasestates, the phase shifters are both set to switch between 0° and 180°,the two classic phase states for BPSK. Consequently, the phase modulatedsignals (Signals 1 and 2, FIG. 10) look like those shown in FIG. 8 if I,Q bit streams are made the same.

As a phase of 0° is identical to its conjugated phase −0°, and similarlya phase of 180° is identical to −180°, Signals 3 and 1 look identical.Due to the additional 180° phase shift experienced by Signal 4, it isthe exact mirror of Signal 2, as seen in FIG. 9.

Signals 1 and 3 are exactly in phase with each other. Hence, oncombination in the first combination phase, Signal 5 will be exactly thesame, only amplified. Signals 2 and 4 on the other hand are exactly inantiphase and will negate each other on combination. Hence, there willbe no Signal 6 present, (see FIG. 10).

Finally, when Signals 5 and 6 are combined, they result in the outputSignal 7 which is identical to Signal 5 due to the lack of Signal 6, asin FIG. 11.

The resulting phase states of the phase modulator are given below. TABLE1 Phase States of the Phase Modulator for BPSK PM_(I) O/P PM_(Q) O/P IPhase State Q = I Phase State 0 180° 0 180° 1  0° 1  0°

2.1.2 Modified Phase State Arrangement

The arrangement described in Section 2.1.1 has the disadvantage that awhole branch is energised but its output is not efficiently used, assignal 2′ and 4′ are cancelled upon recombination. This is at best awaste of available hardware, and if the branch cannot be turned off, itwill continue to consume energy and hence reduce transmitter efficiency.

The initial phase states can be chosen arbitrarily. This feature can beused to set the output power level from the system without having toadjust the amplifiers away from their maximum power added efficiencyoperating point. For example by choosing 45° for the phase staterepresenting a binary “1”, and −135° to represent a binary “0”respectively, the topology can work with maximum efficiency for thismodulation scheme. This aspect is not discussed further.

2.2 QPSK [1]

The case of QPSK has been dealt with in section 1.1. Here, only thephase table is given: TABLE 2 Phase States of the Phase Modulator forQPSK PM_(I) O/P PM_(I) O/P I Phase State Q Phase State 0 135° 0 −45° 1 45° 1 135°

2.3 OFFSET-QPSK or OQPSK [11]

OQPSK is a special case of QPSK. The phase state diagram is exactlyidentical to QPSK, but the I and Q signal are never allowed to changesimultaneously, i.e the difference between the two modulation schemes isonly visible in the time domain. To prevent the I and Q signal fromchanging simultaneously, both are allocated different timeslots in whichthey are allowed to vary, so that even if both I and Q are required tochange, they do so after each other. This way the output vector willnever pass through the origin of the phase diagram, meaning that theoutput signal will never drop to zero. As in every other respect, OQPSKis completely identical with QPSK, all the derivations made in therelevant sections are applicable to OQPSK. TABLE 3 Phase States of thePhase Modulator for OQPSK PM_(I) O/P PM_(Q) O/P I Phase State Q PhaseState 0 135° 0 −45° 1  45° 1 135°

2.4 THE HIGHER-ORDER DIGITAL MODULATION SCHEMES 2.4.1 Relationshipbetween Input and Output Phase

Because of the major number of possible phase states, 8-PSK is no longerdealt with by showing the diagrams for each case, but instead a generalformula for the input phases in dependence of the output is derived, anda table of all phase states created using this formula. The same formulawill also be used to derive the phase state tables in all of thefollowing sections.

To start, the required output RF Signal, Signal 7, V_(RF) in FIG. 12.The phase angle, φ_(RP), can take any value out of range from −180° to+180°, with VRP being the amplitude of the RF signal.

The RF signal is made up of two signals that are orthogonal to eachother, but which in general can have different amplitude, namely Signal5 and Signal 6. Working back from the required amplitude and phaserelationships required for V_(RF), their amplitudes can be derivedthrough trigonometric calculations as:V ₅ =V _(RF) cos (φ_(RF))   (1)V ₆ =V _(RF) sin (φ_(RF))   (2)

Signals V₅ and V₆ again are themselves created through the summation ofphase conjugated signal pairs, namely Signals 1 and 3 in case of V₅, andSignals 2 and 4 in case of V₆. In addition to being phase conjugated inthe arrangement in FIG. 1 (used here for discussion as a means ofrealisation), Signals 2, 4 also receive a 180° phase shift with respectto the LO (see FIG. 1). Next, since Signals 1 and 3 as well as Signals 2and 4 are phase conjugated, and V1=V3, V2=V4, then φ₁=−φ₃, and finally180°−φ₂=−φ₄, respectively, is used to derive the output signals afterthe phase modulators PM1 and PM2:V ₅=2·g·V ₁·cos (φ₁)   (3)V ₆=2·g·V ₂·sin (φ₂)   (4)

Where g is the total gain provided by each amplifier pair.

Equation (1) is inserted in equation (3), and equation (2) is insertedin equation (4), respectively. The resulting equations are solved forthe required phases φ₁ and φ₂ required at PM₁ and PM_(Q) respectively,FIG. 1 obtained. These represent the wanted relationship betweennecessary input signal phases with respect of the desired RF signalphase needed to synthesise any particular digital modulation scheme.$\begin{matrix}{\varphi_{1} = {- {\arccos\left( \frac{V_{RF} \cdot {\cos\left( \varphi_{RF} \right)}}{2 \cdot g \cdot V_{1}} \right)}}} & (5) \\{\varphi_{2} = {- {\arcsin\left( \frac{V_{RF} \cdot {\sin\left( \varphi_{RF} \right)}}{2 \cdot g \cdot V_{2}} \right)}}} & (6)\end{matrix}$

These equations can now be used to set up the tables of phase states forthe higher-order modulation schemes.

2.4.2 8PSK[11]

An 8 PSK signal consists of tribits, i.e. each word is made up of threebits, I,Q, and C. Hence, the signal takes one out of 8 possible phasestates, each with a constant amplitude. Table 4 has been created usingthe equations (5) and (6), and under the assumptions of$V_{1,2} = {\frac{V_{RF}}{2 \cdot g}.}$

It shows the output phase φ_(RF), and the two required input phases, φ₁and φ₂, for each of the input bits combinations. To summarise the table,both phase modulators, PM₁ and PM_(Q) have to create the same phase asrequired for the output signal. TABLE 4 8psk Phase States I Q C φ_(RF)φ₁ φ₂ 0 0 0 −112.5° −112.5° −112.5° 0 0 1 −157.5° −157.5° −157.5° 0 1 0−67.5° −67.5° −67.5° 0 1 1 −22.5° −22.5° −22.5° 1 0 0 +112.5° +112.5°+112.5° 1 0 1 +157.5° +157.5° +157.5° 1 1 0 +67.5° +67.5° +67.5° 1 1 1+22.5° +22.5° +22.5°

2.4.3 16 PSK [11]

For 16 PSK each word consists of four bits, I, Q, C₁ and C₂. Hence, thesignal takes one out of 16 possible phase states, but has constantamplitude. The following table has been created using the equations (5)and (6), and under the assumptions of$V_{1,2} = {\frac{V_{RF}}{2 \cdot g}.}$

It shows the output phase φ_(RF), and the two input phases, φ₁ and φ₂,for each of the input bits comminations. To summarize the table, bothphase modulators, PM₁ and PM_(Q) have to create the same phase asrequired for the output signal. I Q C1 C2 φ_(RF) φ₁ φ₂ 0 0 0 0 11.25°11.25° 11.25° 0 0 0 1 33.75° 33.75° 33.75° 0 0 1 0 56.25° 56.25° 56.25°0 0 1 1 78.75° 78.75° 78.78° 0 1 0 0 101.25° 101.25° 101.25° 0 1 0 1123.75° 123.75° 123.75° 0 1 1 0 146.25° 146.25° 146.25° 0 1 1 1 168.75°168.75° 168.75° 1 0 0 0 −168.75° −168.75° −168.75° 1 0 0 1 −146.25°−146.25° −146.25° 1 0 1 0 −123.75° −123.75° −123.75° 1 0 1 1 −101.25°−101.25° −101.25° 1 1 0 0 −78.75° −78.75° −78.75° 1 1 0 1 −56.25°−56.25° −56.25° 1 1 1 0 −33.75° −33.75° −33.75° 1 1 1 1 −11.25° −11.25°−11.25°

2.4.4 8 QAM [11]

An 8 QAM signal consists of tribits, i.e word consisting of three bits,I, Q, and C. The signal takes one out of 4 possible phase states, whilethe last bit, C, determines the amplitude. The following table has beencreated using the equations (5) and (6), and under the assumptions of$V_{1,2} = \frac{V_{RF}}{2 \cdot g}$for C=1. For C=0, the output amplitude V_(RF,O) is only 0.41 V_(RF).

Hence, the calculation is based on the definition$V_{1,2} = {0.2 \cdot {\frac{V_{RF}}{2 \cdot g}.}}$

Since for QAM signal not all phase state vectors have the same length,those with reduced input amplitude are synthesised purely by choosingcertain input phase angles, φ₁ and φ₂; such that vector combination ofequilength vectors at 5, 6 and subsequently at 7 in FIG. 1 lead to thecorrect phase/magnitude for a prescribed bit pattern. Therefore, achange of input signal amplitude is not necessary. The table shows theoutput amplitude, V_(RF), and output phase φ_(RF), and the two requiredinput phases, φ₁ and φ₂, for each of the input bits combinations. TABLE5 8QAM Phase States I Q C V_(FR)/g′V¹ φ_(RF) φ₁ φ₂ 0 0 0 0.41 −135°   98.4°  −8.4° 0 0 1 1 −135° −135° −135° 0 1 0 0.41 −45°    81.6° −8.4° 0 1 1 1 −45°  −45°  −45° 1 0 0 0.41 135°    98.4°  −8.4° 1 0 1 1135°   135°   135° 1 1 0 0.41 45°    81.6°    8.4° 1 1 1 1 45°    45°   45°

2.4.5 16 QAM [11]

For an 16 QAM signal each word consisting of four bits, namely I, Q, C1,and C2. The signal takes one out of 9 possible phase states, and one ofthree different possible amplitudes. The following table has beencreated using the equations (5) and (6), and under the assumptions of$V_{1,2} = \frac{V_{RF}}{2 \cdot g}$for the largest output signal amplitude,$V_{2} = {0.37\quad{\frac{V_{RF}}{2 \cdot g}.}}$for the medium amplitude, and$V_{1,2} = {0.13 \cdot {\frac{V_{RF}}{2 \cdot g}.}}$

for the small amplitude. As before for 8 QAM, the reduced input areachieved purely by choosing certain input phase angles, φ₁ and φ₂, whichwhen vector added lead to the required amplitude value such that achange of input signal amplitude is not necessary. The table shows theoutput amplitude, V_(RF), and output phase φ_(RF), and the two requiredinput phases, φ₁ and φ₂, for each of the input bits combinations. I Q C₁C₂ V_(RF)/g · V_(1,2) φ_(RF) φ₁ φ₂ 0 0 0 0 0.52 −135° 100.6° −10.6° 0 00 1 1.46 −165° 134.8° −10.9° 0 0 1 0 0.52 −45°  79.4° −10.6° 0 0 1 11.46 −15°  45.2° −10.9° 0 1 0 0 1.46 −105° 100.9° −44.8° 0 1 0 1 2 −135°135° −45° 0 1 0 0 1.46 −75°  79.1° −44.8° 0 1 1 1 2 −45°  45° −45° 1 0 10 0.52 135° 100.6°   10.6° 1 0 0 1 1.46 175° 136.7°    3.6° 1 0 0 0 0.5245°  79.4°   10.6° 1 0 1 1 1.46 15°  45.2°   10.6° 1 1 1 0 1.46 105°100.9°   44.8° 1 1 0 1 2 135° 135°   45° 1 1 1 0 1.46 75°  79.10°  44.8° 1 1 1 1 2 45°  45°   45°

2.5 AM/FM MODULATION

If the basic phase conjugation technique described in Section 1.1 isapplied to FIG. 13(a), where a(t) is an analogue input signal and PM isan analogue phase shifter that can provide up to +90° phase shift, thenthe arrangement can be made to produce an amplitude modulated (AM) RFcarrier, while that in FIG. 13(b) will produce Frequency Modulation (FM)by the direct FM method [11].

3 PROOF OF CONCEPT

Simulations were carried out using Hewlett Packard's Advanced DesignSystem (ADS) [6] to simulate the circuit at a systems level; i.e.building blocks for mixers, phase shifters etc. were used to simulateideal circuit components. The results confirmed that all the high-levelmodulation digital schemes listed above can be produced using theschemes described.

The arrangement in FIG. 1 was realised using commercially availablecomponents. FIGS. 14 a and 14 b show the measured phase states for QPSKand 8 QAM operation. The maximum output power deviation is 0.47 dB, andthe maximum phase error 3°.

Simulation was also used to investigate the effects of non-linearamplification. The model used was an RFIC IQ Modulator power amplifier,taken from the ADS example library. This model allows for most of themajor dominant factors that reduce linearity to be taken into account,e.g. third order intercept point, gain saturation, harmonic generation,etc.

FIG. 14 is a simulated phase diagram for a QPSK system using the schemeof FIG. 1, with linear amplification. FIG. 15 shows the equivalent phasediagram for the non-linear simulation. It can be seen that there are twomain effects due to the inclusion of the non-linear PA. One is thegeneration of harmonics, which can be seen in the centre of the polardiagram, while the group delay of the PA has turned the whole diagramaround the centre. The latter effect is not really a problem, as usuallythe receiver does not require an absolute phase reference for signalrecovery. In practice, any phase shifts introduced by the PAs can beaccounted for by introducing equalizing phase shift, e.g. by means of atransmission line placed immediately in front of each of the amplifiersin FIG. 1, FIG. 6, FIG. 7.

FIG. 16 shows the spectra on four different positions in the circuit.The uppermost spectrum is taken directly before the PA, showing thespectral clarity of the input signal with all harmonics about 100 dBbelow carrier (signal 1 in FIG. 1). The second plot shows the spectrumdirectly after the PA, which has driven into saturation during thesimulation resulting in a significant number of strong harmonics (signal1′ in FIG. 1). The third diagram shows the combined signals 1′ and 3′from the upper branch of the system (signal 5 in FIG. 1). It can be seenthat every second even harmonic, starting with the second one, i.e n=2,6, 10 . . . , has been cancelled out upon recombination. The lowerdiagram finally shows the spectrum that results when signal 1′ issubtracted from the signal 3′ rather than combined with it. It can beseen that again every second even harmonic starting with the 4^(th) one,i.e n=4, 8, 12 . . . , has been cancelled. This cancellation effect uponsubtraction is in line with that predicted by Cox [2].

As can be seen, by adding the signals, some of the harmonics includingthe second one are balanced out. Now given that the third harmonic wouldtypically be outside the bandwidth of most antennas, little or even noRF filtering would be required for the given system.

Thus, it is seen that strong harmonic production in the amplifiers due(in this case) to their forced working in a non-linear regime does notsignificantly affect the overall linear output performance of thesystem. This opens up the way for the use of highly efficient highlynon-linear (i.e. DC to RF efficient) amplifiers such as a class E (85%efficiency) [7] in the system.

4 COMPARISON WITH PRIOR ART 4.1 CLASSIC LINEAR TRANSMITTER TOPOLOGIES4.1.1 Superheterodyne Transmitters

In modern communications systems that require a high degree oflinearity, modulation usually takes place at a low frequency, and oncemodulated, the signal is up converted to the RF frequency then amplifiedusing a linearised amplifier. Since with this strategy the amplifiershave to be highly linear, they have an inherently low DC to RFefficiency which makes them inefficient for mobile communicationsapplications.

A review of the efficient linear transmitter schemes (all withoutintegrated modulators) is given in [8]. These are briefly summarizedbelow to better set our work in context.

4.1.2 Feedback Transmitter

These are linear amplifier structures where a small part of the radiatedsignal is demodulated locally the transmitter. In a second step,information about the amplifier's non-linearities gained from thisdemodulated control signal is then used to modify the input signaland/or make corrections to the settings of the power amplifier sectionin order to increase linearity of the power amplification. Two popularmethods are the Cartesian feedback loop, and the adaptive pre-distortionamplifier [8].

While the feedback systems have shown the potential to use amplifierswith significantly reduced linearity, and hence high DC to RFefficiency, they however all suffer from one inherent disadvantage,which is the inevitable feedback loop with high loop gain, thisarrangement makes this class of power amplifiers highly susceptible tooscillations [8].

The system of the invention does not require feedback for its operation,also, unlike the above, modulator and transmitter functions are combinedand are digital modulation scheme independent.

4.1.3 LINC Structures

The idea here is to use two highly efficient amplifiers each amplifyinga constant envelope phase modulated signal, which are subsequentlycombined to give an AM, or a PM signal. First introduced by Cox in 1974[2], his initial idea suffered from a complexity of realisation, andsoon further advances were made, e.g. [9], [10]. The current approach isto obtain the required phase conjugated signals through DSP methods toavoid the problems of the complex LINC hardware structure proposed byCox [2]. This approach however is limited to low data rates due to DSPprocessor limitations.

While the work presented here follows the same basic principle of usingconstant amplitude signals during amplification, there are howeversignificant differences in that first we do not produce AM and PMsignals, second the modulation process takes place at the RF frequency,making the up-converters redundant, and third the inherent properties ofeither a mixer or a See-Saw modulator are used to create the phaseconjugated signals, hence removing the need for large “signalseparation” circuitry that was required by Cox. Furthermore, by usingtwo separate amplifier chains in quadrature to each other, we gain thepossibility to create any given modulation scheme just by choosingdifferent phase angles for two phase modulators. This aspect has notbeen addressed previously. The problem of integral up conversion andadaptive modulation have to date been partly addressed, [12]. Thearchitecture in [12] represents a very complex solution and isconsiderably less flexible than the one suggested here, since itrequires the use of injection locked oscillators and a complex matrixswitch arrangement for its operation.

With our system, in order to create any of the above given modulationschemes, only the phase angles of the input phase shifters have to bepreset to a finite number of possible phase states. Consequently, thissuggested topology allows for the first time the creation of a truedigital multimode modulator/transmitter that can be switched between anyof the commonly used data communication modulation schemes withouthaving sophisticated hardware or DSP incorporated.

Additionally, a second technique using a See-Saw modulator wasintroduced. In this case, shorted lines switched by PIN diodes are usedto create the necessary phase shifts and conjugated signals. While thistechnique lacks the total flexibility of the initial concept, itnevertheless introduced a very simple way to create the necessary phasemodulated signals without the need of doublers or mixers and would beexceptionally useful in all wireless frequency bands, especially thoseassociated with millimetre-wave broadband wireless systems where thearrangement suggested could easily be implemented in MMIC form.

4.2 Modulators

The state-of-the-art approach to any of the above mentioned modulationschemes is to have a baseband modulator, which is tailor-made for thespecific modulation scheme in question. The baseband signal is then upconverted to the RF frequency, linearly amplified, and transmitted.While it is known that in principle a linear IQ modulator [11] cancreate any output phase and amplitude, such a universal device embodyinginherent up conversion has not yet been realised. Therefore a freelydefinable IQ modulator/high frequency transmitter was until now thoughtto be too complicated to be of practical value.

In contrast, the approach in our work uses simple phase shifters tocreate the correct phase vector component used to synthesise the digitalmodulation states directly at the RF carrier frequency. This eliminatesthe need for D/A converters and sophisticated DSP, and keeps themodulators simple enough to work directly at the RF band, thereforeremoving the need for up conversion required by classical systems.

4.3 Parallel Amplification

In the present system, because the in-phase and the quadrature part ofthe signal are kept separate until the very last stage, they canactually be amplified independently prior to combination. The system canif required be made maximally efficient by including a power factorcorrection feature into the combination circuitry. This means that eachof the four power amplifiers in each of the four signal paths have toachieve significantly less output power than a single power amplifierwould have to achieve in a classic, single path topology. As it is mucheasier to build several power amplifiers for relatively low power levelsthan building a single amplifier for high power, this is a significantadvantage of the proposed system over classic approaches.

In addition, these amplifiers can be highly nonlinear without impairingoverall quality of the modulation produced.

For some applications, the classic power amplifier cannot be built witha single device because the device cannot handle the currents requiredto achieve the desired output power. Many classic amplifiers willactually consist of several devices used in parallel. In addition to therequired splitters, additional control circuitry is required foramplifier linearisation.

However, there is one disadvantage of our proposed system over a classicpower amplifier, which is that for high order modulation schemes themaximum RF output power is not twice the output power of one branch ofthe system. The reason is that the most efficient way to combine signalswould be to combine them in-phase, not in quadrature. Especially, forsignals that have a phase angle which is close to a multiple of 90°,either the in-phase or the quadrature path contributes very little tothe overall output power which is left to the other path. Thisphenomenon is smallest for QPSK or BPSK, and becomes more dominant forthe higher the modulation order schemes, i.e. worst for 16 PSK and 16QAM. However, this phenomenon is likely to be compensated for by thefact that the requirements for the linearity of the power amplifiers ismade redundant thereby allowing for them to be driven very hard and tooperate very effectively.

4.4 Linearisation Requirements for QAM

Classic solutions for QAM signals have to be highly linear, as theamplitude of the signal contains modulation information. This putsparticularly critical requirements on the final power amplifier, whichhas to achieve high output power levels and high linearity, whilesimultaneously being energy efficient to secure sufficient battery lifefor mobile unit operation. Unfortunately, the most energy efficientoperating modes for power amplifier are also the most non-linear ones,and hence much more suitable to constant envelope signals. Due to theserestrictions, classic amplifier solutions always have to compromiseefficiently for linearity to some degree.

In the system proposed here, the in-phase and quadrature signals areeach phase modulated and consequently have constant envelope duringamplification. Hence, the power amplifiers for the system suggested herecan work in their highly efficient non-linear region withoutcompromising signal integrity.

Additionally, as predicted by Cox [2], some of the harmonics generatedin each of the branches are in antiphase, and others in phase, withrespect to the other branch. As a result, when the signals are combined,cancellation takes place, actively increasing the linearity of thesystem.

4.5 Summary

Systems embodying the invention have the following advantages overclassic topologies:

-   -   High flexibility; any of the commonly used digital modulation        schemes can be implemented without changing the hardware, thus        yielding a major step towards a software-reconfigurable digital        radio.    -   The core of the system can be easily modified for analogue AM or        FM modulation.    -   Constant envelope signal during amplification phases results in        -   High linearity        -   Simple nonlinear amplifiers        -   High DC to RF efficiency        -   Inherent cancellation of some of the harmonics, i.e., the            system is actively increasing linearity        -   No feedback loop removes risk of oscillations        -   Low component count compared to standard LINC technology. No            DSP, hence no sample rate and aliasing problems, or A/D            converters needed, and enhanced flexibility in types of            modulation schemes produced            References

-   [1] P. B. Kenington: “Emerging technologies for software radio”, IEE    Electronics & Communication Engineering Journal, April 1999, pp.    69-83

-   [2] D. C. Cox: “Linear Amplification with Nonlinear Components”,    IEEE Transactions on Communications, December 1974, pp 1942-1945

-   [3] S. A. Maas: “The RF and Microwave Circuit Design Cookbook”,    Artech House, Norwood, 1998, pp 236-240

-   [4] J. A. Navarro, K Chang: “Integrated Active Antennas and Spatial    Power Combining”, Wiley, 1996

-   [5] S. Foti, R Cahill, C. Kaniou: “See Saw phase shifter cuts cost    of phased array”, Microwaves & RF, March 1987, pp69-77

-   [6] “Advanced Design System”, Version 1.5, Agilent Technologies,    2000

-   [7] F Raab: “Class-E, Class-C and Class-F power amplifiers based    upon a finite number of harmonics”, IEEE Trans. MTT, Vol 49, No 8,    August 2001, pp 1462-1468

-   [8] S. Mann, M. Beach, P. War, J McGeechan: “Increasing the    Talk-Time of Mobile Radios with efficient linear Transmitter    Architectures”, Electronics and Communications Engineering Journal,    April 2001, pp 65-76

-   [9] A. Bateman, D. M. Haines, and R. J. Wilkinson: “Linear    Transceiver Architectures”, IEEE Vehicular Technology Conf., May    1988, pp 478-484

-   [10] S. A. Hetzel, A. Bateman, and J P McGeechan: “LINC    Transmitter”, Electronics Letters, May 1991, Vol 27, No 10, pp    844-846

-   [11] W. Tomasi: “Electronic Communications Systems”, Prentice Hall,    Inc., Englewood Cliffs, 1988

-   [12] D J Jennings, J P McGreehan: “A high-Efficiency RF Transmitter    using VCO-derived Synthesis: CALLUM, IEEE Trans. MTT, Vol 47, No 6,    June 1999, pp 715-721

1. A method for providing an amplified modulated radio frequency signal,the method comprising: providing a radio frequency (RF) oscillatorysignal; generating from the RF oscillatory signal and an input signal apair of phase modulated phase conjugated signals; and summing said pairof signals to provide the desired modulated RF signal output.
 2. Themethod of claim 1, wherein said pair of signals may be produced by:modulating said RF oscillatory signal with the input signal to provide amodulated RF signal, and combining the modulated RF signal with amultiple of the RF oscillatory signal to provide a composite signal. 3.The method of claim 2, wherein each of the modulated RF signal and thecomposite RF signal is separately amplified before being combined withthe other.
 4. The method of claim 3, wherein the amplification isnon-linear.
 5. The method of claim 2, wherein said multiple is
 2. 6. Themethod of claim 2, wherein said multiple is produced from theoscillatory signal by a frequency multiplier or by use of a harmonicmixer.
 7. The method of claim 2, wherein the input signal is an analoguesignal and said modulation is an analogue phase shift.
 8. The method ofclaim 7, wherein the result of the summation is an amplitude modulatedRF signal.
 9. The method of claim 8, wherein the amplitude modulatedsignal may be further summed with said RF oscillatory signal to providea frequency modulated RF signal.
 10. The method of claim 1, wherein theinput signal comprises in-phase (I) and quadrature (Q) input signalswhich are digitally modulated.
 11. The method of claim 10, wherein eachof the I and Q signals is processed in a respective channel.
 12. Themethod of claim 11, wherein the channel outputs are summed.
 13. Themethod claim 11, wherein the channels share an RF oscillatory signal,which is phase shifted by 90 degrees within the Q channel.
 14. Amodulator for generating a modulated RF signal, the modulatorcomprising: a local oscillator generating an RF oscillatory signal;means for producing, from said RF oscillatory signal and an inputsignal, a pair of phase modulated phase conjugated signals; and asumming circuit receiving said pair of signals as inputs to generate amodulated RF output signal as its output.
 15. The modulator of claim 14,wherein said means for producing a pair of phase modulated phaseconjugated signals comprises: a modulator arranged to modulate said RFoscillatory signal with the input signal to provide a modulated RFsignal, and a mixer connected to mix said modulated RF signal with amultiple of the RF oscillatory signal to provide a composite signal. 16.The modulator of claim 15, wherein a first amplifier is connectedbetween the mixer output and a respective input of a first summer, and asecond amplifier is connected between the modulator output and arespective input of a first summer.
 17. The modulator of claim 15,wherein the mixer is a fundamental mixer and receives the localoscillator output via a multiplier circuit.
 18. The modulator of claim17, wherein the multiplier circuit comprises a doubler.
 19. Themodulator of claim 16, wherein the mixer is a harmonic mixer connecteddirectly to the local oscillator output.
 20. The modulator of claim 14,wherein the input signal is an analogue signal and the modulator is ananalogue phase shifter.
 21. The modulator of claim 20, wherein theoutput of the summing circuit is an amplitude modulated RF signal. 22.The modulator of claim 21, wherein the modulator includes a furthersummer which sums the output of the first summer with the localoscillator signal to provide a frequency modulated output signal. 23.The modulator of claim 14, wherein the input signal comprises in-phase(I) and quadrature (Q) input signals, which are digitally modulated. 24.The modulator of claim 23, wherein the modulator comprises an I channeland a Q channel, each channel sharing a common local oscillator.
 25. Themodulator of claim 24, wherein the outputs of the summers of the I and Qchannels are combined by a third summer to provide the modulated RFoutput signal.
 26. The modulator of claim 24, wherein each channelcomprises a mixer.
 27. The modulator of claim 26, wherein the mixers arefundamental mixers supplied with a frequency doubled local oscillatorsignal.
 28. The modulator of claim 26, wherein each channel has its ownfrequency doubler.
 29. The modulator of claim 28, wherein the frequencydoubler in the Q channel receives the local oscillator output phaseshifted by 90 degrees.
 30. The modulator of claim 26, wherein bothmixers are supplied by a single frequency doubler.
 31. The modulator ofclaim 30, wherein the signal from the local oscillator to the Q channelmodulator is phase shifted by 90 degrees.
 32. The modulator of claim 26,wherein the mixers are harmonic mixers.
 33. The modulator of claim 14,wherein the input signal is provided in a plurality of channels, and themeans for producing the pair of phase modulated phase conjugated signalscomprises a see-saw modulator in each channel.
 34. The modulator ofclaim 33, wherein the see-saw modulator uses shorted lines to createphase shifts for a phase modulated signal.
 35. The modulator of claim34, wherein the lines are selectively shorted electronically.
 36. Themodulator of claim 35, wherein the lines are shorted by switching of PINdiodes.